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Low Power, Low Noise and Distortion, Rail-to-Rail Output Amplifier ADA4841-1
FEATURES
Low power: 1.1 mA supply current Low wideband noise 2 nV/Hz 1.4 pA/Hz Low 1/F noise 6 nV/Hz @ 10 Hz 13 pA/Hz @ 10 Hz Low distortion -111 dBc @ 100 kHz, VO = 2 V p-p High speed 80 MHz, -3 dB bandwidth (G = +1) 12 V/s slew rate 175 ns settling time to 0.1% Low offset voltage 0.29 mV max Rail-to-rail output Power down Wide supply range: 2.7 V to 12 V
CONNECTION DIAGRAM (TOP VIEW)
NC 1 -IN 2 +IN 3 -VS 4 8 POWER DOWN 7 +VS
05614-001
6 VOUT 5 NC
Figure 1. 8-Lead SOIC (R)
APPLICATIONS
Low power, low noise signal processing Battery-powered instrumentation 16-bit PulSAR(R) ADC drivers
GENERAL DESCRIPTION
The ADA4841-1 is a unity gain stable, low noise and distortion, rail-to-rail output amplifier that has a quiescent current of 1.4 mA maximum. Despite its low power consumption, this amplifier offers low wideband voltage noise performance of 2 nV/Hz and 1.4 pA/Hz current noise, along with excellent spurious-free dynamic range (SFDR) of -110 dBc at 100 kHz. To maintain a low noise environment at lower frequencies, the amplifier has low 1/F noise of 6 nV/Hz and 13 pA/Hz at 10 Hz. The ADA4841-1 has a wide supply voltage range from 2.7 V to 12 V and an output that swings within less than 100 mV of either rail. The input common-mode voltage range extends down to the negative supply. The ADA4841-1 can drive up to 20 pF of capacitive load with minimal peaking. The ADA4841-1 provides the performance required to efficiently support emerging 16-bit to 18-bit ADCs and is ideal for portable instrumentation, high channel count, industrial measurement, and medical applications. The ADA4841-1 is an ideal match for driving Analog Devices, Inc. AD7685/AD7686 16-bit PulSAR ADCs.
Rev. A
Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners.
The ADA4841-1 is a SOIC package, with Pb-free lead finish. This amplifier is rated to work over the industrial temperature range (-40C to +125C).
-30 -40 VS = 5V G = +1
HARMONIC DISTORTION (dBc)
-50 -60 -70 2V p-p THIRD -80 -90 -100 -110 -120 0.01 2V p-p SECOND
05614-048
0.1 FREQUENCY (MHz)
1
Figure 2. Harmonic Distortion
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781.329.4700 www.analog.com Fax: 781.461.3113 (c) 2005 Analog Devices, Inc. All rights reserved.
ADA4841-1 TABLE OF CONTENTS
Features .............................................................................................. 1 Applications....................................................................................... 1 Connection Diagram (Top View)................................................... 1 General Description ......................................................................... 1 Revision History ............................................................................... 2 Specifications..................................................................................... 3 Absolute Maximum Ratings............................................................ 6 Thermal Resistance ...................................................................... 6 ESD Caution.................................................................................. 6 Typical Performance Characteristics ............................................. 7 Theory of Operation ...................................................................... 13 Amplifier Description................................................................ 13 DC Performance Considerations ............................................. 13 Noise Considerations ................................................................. 13 Headroom Considerations........................................................ 14 Capacitance Drive ...................................................................... 15 Input Protection ......................................................................... 15 Power-Down Operation ............................................................ 16 Applications..................................................................................... 17 Typical Performance Values...................................................... 17 16-Bit ADC Driver..................................................................... 17 Reconstruction Filter ................................................................. 17 Layout Considerations............................................................... 18 Ground Plane.............................................................................. 18 Power Supply Bypassing ............................................................ 18 Outline Dimensions ....................................................................... 19 Ordering Guide............................................................................... 19
REVISION HISTORY
9/05--Rev. 0 to Rev. A Changes to Features.......................................................................... 1 Changes to Figure 2.......................................................................... 1 Changes to Figure 12........................................................................ 8 Changes to Figure 40...................................................................... 14 Changes to Headroom Considerations Section ......................... 15 7/05--Revision 0: Initial Version
Rev. A | Page 2 of 20
ADA4841-1 SPECIFICATIONS
TA = 25C, VS = 5 V, RL = 1 k, Gain = +1, unless otherwise noted. Table 1.
Parameter DYNAMIC PERFORMANCE -3 dB Bandwidth Slew Rate Settling Time to 0.1% Settling Time to 0.01% NOISE/HARMONIC PERFORMANCE Harmonic Distortion HD2/HD3 Input Voltage Noise Input Current Noise DC PERFORMANCE Input Offset Voltage Input Offset Voltage Drift Input Bias Current Input Offset Current Open-Loop Gain INPUT CHARACTERISTICS Input Resistance, Common Mode Input Resistance, Differential Mode Input Capacitance, Common Mode Input Capacitance, Differential Mode Input Common-Mode Voltage Range Common-Mode Rejection Ratio (CMRR) POWER DOWN Input Voltage Power Down Input Bias Current Enable Power Down Switching Speed Enable Power Down OUTPUT CHARACTERISTICS Output Voltage Swing Output Current Capacitive Load Drive POWER SUPPLY Operating Range Quiescent Current Positive Power Supply Rejection Ratio Negative Power Supply Rejection Ratio Conditions VO = 0.02 V p-p VO = 2 V p-p G = +1, VO = 9 V step, RL = 1 k G = +1, VO = 8 V step G = +1, VO = 8 V step fC = 100 kHz, VO = 2 V p-p, G = +1 fC = 1 MHz, VO = 2 V p-p f = 100 kHz f = 100 kHz Min 58 12 Typ 80 3 13 650 1000 -111/-115 -80/-67 2 1.3 60 1 3.4 0.1 114 90 25 1 3 VCM = -2 V to +2 V -5.1 95 +4 114 290 5.3 0.4 Max Unit MHz MHz V/s ns ns dBc dBc nV/Hz pA/Hz V V/C A A dB M k pF pF V dB
VO = 4 V
103
3.3 Power down = +5 V Power down = -5 V 1 -13 1 40 4.90 Sourcing Sinking 30% overshoot 2.7 Disable = Low +VS = 5 V to 6 V, -VS = -5 V +VS = 5 V, -VS = -5 V to +6 V 1.1 40 110 120 4.955 30 60 15 12 1.5 90 2 -30
V A A s s V mA mA pF V mA A dB dB
96 97
Rev. A | Page 3 of 20
ADA4841-1
TA = 25C, VS = 5 V, RL = 1 k, Gain = +1, unless otherwise noted. Table 2.
Parameter DYNAMIC PERFORMANCE -3 dB Bandwidth Slew Rate Settling Time to 0.1% Settling Time to 0.01% NOISE/HARMONIC PERFORMANCE Harmonic Distortion HD2/HD3 Input Voltage Noise Input Current Noise DC PERFORMANCE Input Offset Voltage Input Offset Voltage Drift Input Bias Current Input Offset Current Open-Loop Gain INPUT CHARACTERISTICS Input Resistance, Common Mode Input Resistance, Differential Mode Input Capacitance, Common Mode Input Capacitance, Differential Mode Input Common-Mode Voltage Range Common-Mode Rejection Ratio (CMRR) Power Down Input Voltage Power Down Input Bias Current Enable Power Down Switching Speed Enable Power Down OUTPUT CHARACTERISTICS Output Voltage Swing Output Current Capacitive Load Drive POWER SUPPLY Operating Range Quiescent Current Positive Power Supply Rejection Ratio Negative Power Supply Rejection Ratio Conditions VO = 0.02 V p-p VO = 2 V p-p G = +1, VO = 4 V step, RL = 1 k G = +1, VO = 2 V step G = +1, VO = 2 V step fC = 100 kHz, VO = 2 V p-p fC = 1 MHz, VO = 2 V p-p f = 100 kHz f = 100 kHz Min 54 10 Typ 80 3 12 175 550 -109/-114 -78/-66 2 1.3 60 1 3.4 0.1 124 90 25 1 3 VCM = 0 V to1.5 V -0.1 88 +4 120 290 5.3 0.4 Max Unit MHz MHz V/s ns ns dBc dBc nV/Hz pA/Hz V V/C A A dB M k pF pF V dB
VO = 0.5 V to 4.5 V
103
3.3 Power down = 5 V Power down = 0 V 1 -13 1 40 0.065 to 4.920 Sourcing Sinking 30% overshoot 2.7 Power down = 0 V +VS = 5 V to 6 V, -VS = 0 V +VS = 5 V, -VS = 0 V to -1 V 1.1 35 110 120 0.029 to 4.974 30 60 15 5.5 1.4 70 2 -30
V A A s s V mA mA pF V mA A dB dB
95 96
Rev. A | Page 4 of 20
ADA4841-1
TA = 25C, VS = 3 V, RL = 1 k, Gain =+1, unless otherwise noted. Table 3.
Parameter DYNAMIC PERFORMANCE -3 dB Bandwidth Slew Rate Settling Time to 0.1% Settling Time to 0.01% NOISE/HARMONIC PERFORMANCE Harmonic Distortion HD2/HD3 Input Voltage Noise Input Current Noise DC PERFORMANCE Input Offset Voltage Input Offset Voltage Drift Input Bias Current Input Offset Current Open-Loop Gain INPUT CHARACTERISTICS Input Resistance, Common Mode Input Resistance, Differential Mode Input Capacitance, Common Mode Input Capacitance, Differential Mode Input Common-Mode Voltage Range Common-Mode Rejection Ratio (CMRR) Power Down Input Voltage Power Down Input Bias Current Enable Power Down Switching Speed Enable Power Down OUTPUT CHARACTERISTICS Output Voltage Swing Output Current Capacitive Load Drive POWER SUPPLY Operating Range Quiescent Current Positive Power Supply Rejection Ratio Negative Power Supply Rejection Ratio Conditions VO = 0.02 V p-p G = +1, VO = 2 V step, RL = 1 k G = +1, VO = 1 V step G = +1, VO = 1 V step fC = 100 kHz, VO = 1 V p-p fC = 1 MHz, VO = 1 V p-p f = 100 kHz f = 100 kHz Min 52 10 Typ 80 12 120 250 -97/-100 -79/-80 2 1.3 60 1 3.4 0.1 123 90 25 1 3 VCM = 0 V to 0.4 V -0.1 86 +2 120 295 5.3 0.4 Max Unit MHz V/s ns ns dBc dBc nV/Hz pA/Hz V V/C A A dB M k pF pF V dB
VO = 0.5 V to 4.5 V
101
1.3 Power down = 3 V Power down = 0 V 1 -10 1 40 0.045 to 2.955 Sourcing Sinking 30% overshoot 2.7 Power down = 0 V +VS = 3 V to 4 V, -VS = 0 V +VS = 3 V, -VS = 0 V to -1 V 1.1 25 110 120 0.023 to 2.988 30 60 30 3.5 1.3 60 2 -30
V A A s s V mA mA pF V mA A dB dB
95 96
Rev. A | Page 5 of 20
ADA4841-1 ABSOLUTE MAXIMUM RATINGS
Table 4.
Parameter Supply Voltage Power Dissipation Common-Mode Input Voltage Differential Input Voltage Storage Temperature Operating Temperature Range Lead Temperature Junction Temperature Rating 12.6 V See Figure 3 -VS - 0.5 V to +VS + 0.5 V 1.8 V -65C to +125C -40C to +85C JEDEC J-STD-20 150C
The power dissipated in the package (PD) is the sum of the quiescent power dissipation and the power dissipated in the die due to the amplifier's drive at the output. The quiescent power is the voltage between the supply pins (VS) times the quiescent current (IS). PD = Quiescent Power + (Total Drive Power - Load Power)
V V V 2 PD = (VS x I S ) + S x OUT - OUT RL RL 2 RMS output voltages should be considered. If RL is referenced to -VS, as in single-supply operation, the total drive power is VS x IOUT. If the rms signal levels are indeterminate, consider the worst case, when VOUT = VS/4 for RL to midsupply.
Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability.
PD = (VS x I S ) +
(VS /4 )2
RL
THERMAL RESISTANCE
JA is specified for the worst-case conditions, that is, JA is specified for device soldered in circuit board for surface-mount packages. Table 5. Thermal Resistance
Package Type 8-lead SOIC JA 125 Unit C/W
In single-supply operation with RL referenced to -VS, worst case is VOUT = VS/2. Airflow increases heat dissipation, effectively reducing JA. In addition, more metal directly in contact with the package leads and through holes under the device reduces JA. Figure 3 shows the maximum safe power dissipation in the package vs. the ambient temperature for the 8-lead SOIC (125C/W) on a JEDEC standard 4-layer board. JA values are approximations.
2.0
Maximum Power Dissipation
The maximum safe power dissipation for the ADA4841-1 is limited by the associated rise in junction temperature (TJ) on the die. At approximately 150C, which is the glass transition temperature, the plastic changes its properties. Even temporarily exceeding this temperature limit may change the stresses that the package exerts on the die, permanently shifting the parametric performance of the amplifiers. Exceeding a junction temperature of 150C for an extended period can result in changes in silicon devices, potentially causing degradation or loss of functionality.
MAXIMUM POWER DISSIPATION (W)
1.5
SOIC 1.0
0.5
05614-061
0 -55 -35 -15 5 25 45 65 85 105 AMBIENT TEMPERATURE (C)
125
Figure 3. Maximum Power Dissipation vs. Temperature for a 4-Layer Board
ESD CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although this product features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality.
Rev. A | Page 6 of 20
ADA4841-1 TYPICAL PERFORMANCE CHARACTERISTICS
RL = 1 k, unless otherwise noted.
3 3 VOUT = 2V pp VS = 5V 0 0 G = +10 G = +2 G = +1 VS = 5V VIN = 20mV p-p G = +1 -40C +25C
NORMALIZED CLOSED-LOOP GAIN (dB)
GAIN (dB)
-3
+125C -3
-6
-6 -9
05614-021 05614-028
-12 0.1
1 FREQUENCY (MHz)
10
-9 0.1
1
10 FREQUENCY (MHz)
100
Figure 4. Large Signal Frequency Response vs. Gain
6 VIN = 20mV p-p G = +1 2 20pF 20pF WITH 100 SNUBBER 1 0 -1
Figure 7. Small Signal Frequency Response vs. Temperature
VIN = 20mV p-p G = +1 VS = +5V
VS = +3V
VS = 5V
3
CLOSED-LOOP GAIN (dB)
GAIN (dB)
05614-026
0 0pF -3
-2 -3 -4
10pF
-6
05614-029
-5 -6 0.1
-9 0.1
1
10 FREQUENCY (MHz)
100
1
10 FREQUENCY (MHz)
100
Figure 5. Small Signal Frequency Response vs. Capacitive Load
3 VIN = 20mV p-p VS = 5V G = -1
Figure 8. Small Signal Frequency Response vs. Supply Voltage
3 VS = 5V G = +1
NORMALIZED CLOSED-LOOP GAIN (dB)
G = +1
10mV p-p
0 G = +10
0
GAIN (dB)
-3
-3
2V p-p
400mV p-p 20mV p-p
-6
-6
-9
05614-027
100mV p-p
05614-014
-12 0.1
1
10 FREQUENCY (MHz)
100
-9 0.1
1
10 FREQUENCY (MHz)
100
Figure 6. Small Signal Frequency Response vs. Gain
Figure 9. Frequency Response for Various VOUT
Rev. A | Page 7 of 20
ADA4841-1
140 120 100 80 60 40 20 0 -20 10 PHASE MAGNITUDE VS = 5V 0 -20
-30 -40
OPEN-LOOP PHASE (Degrees)
VOUT = 2V p-p G = +2 +5V SECOND
OPEN-LOOP GAIN (dB)
-40 -60 -80 -100 -120 -140 -160 100M
HARMONIC DISTORTION (dBc)
-50 -60 -70 -80 -90 -100 -110 -120 5V SECOND -130 0.01 0.1 +3V THIRD +3V SECOND
5V THIRD
05614-047
+5V THIRD
100
1k
10k
100k
1M
10M
05614-042
FREQUENCY (Hz)
1
FREQUENCY (MHz)
Figure 10. Open-Loop Gain and Phase vs. Frequency
Figure 13. Harmonic Distortion vs. Frequency for Various Supplies
-30 -40
VS = + 5V VOUT = 2V p-p
10
HARMONIC DISTORTION (dBc)
-50 -60 -70 -80 -90 -100 -110 -120 -130 0.01 0.1 FREQUENCY (MHz) G = +1 SECOND G = +1 THIRD G = +2 THIRD 1
05614-045
05614-034
G = +5 THIRD
G = +2 SECOND G = +5 SECOND
VOLTAGE NOISE (nV/ Hz)
1 10
100
1k
10k
100k
1M
10M
FREQUENCY (Hz)
Figure 11. Harmonic Distortion vs. Frequency for Various Gains
Figure 14. Voltage Noise vs. Frequency
-30 -40
VS = 5V G = +1 8V p-p SECOND 8V p-p THIRD
100
HARMONIC DISTORTION (dBc)
-60 -70 4V p-p THIRD -80 4V p-p SECOND -90 -100 2V p-p THIRD -110 2V p-p SECOND -120 0.01 0.1 FREQUENCY (MHz) 1
05614-046
CURRENT NOISE (pA/ Hz)
-50
10
1
0.1 10
100
1k
10k
100k
1M
FREQUENCY (Hz)
Figure 12. Harmonic Distortion vs. Frequency for Various Output Voltages
Figure 15. Current Noise vs. Frequency
Rev. A | Page 8 of 20
05614-018
ADA4841-1
55 COUNT = 190 50 x = 0.36V/C = 1.21V/C 45
OUTPUT VOLTAGE (V)
0.25 G = +2 TIME = 50ns/DIV VS = +5V VS = +3V
0.24
NUMBER OF PARTS
40 35 30 25 20 15 10 5 0 -5 -4 -2 0 2 4 6
05614-053
0.23 VS = 5V 0.22
0.21
0.20
05614-033
0.19
OFFSET DRIFT DISTRIBUTION (V/C)
Figure 16. Input Offset Voltage Distribution
10 9 8 G = +1 VS = 5V
Figure 19. Small Signal Transient Response for Various Supplies
0.15 G = +2 VIN = 20mV p-p TIME = 50ns/DIV
0.14
OUTPUT VOLTAGE (V)
NONLINEARITY (V)
7 6 5 4 3 2
0pF 0.13
10pF
0.12
0.11
0 0 1 2 VIN (V) 3 4 5
0.09
Figure 17. Nonlinearity vs. VIN
100 80 60
VOFFSET (V)
Figure 20. Small Signal Transient Response for Various Capacitive Loads
0.26 0.25 0.24
VS = 5
G = +2 TIME = 50ns/DIV
40 20 0 -20
05614-036
OUTPUT VOLTAGE (V)
0.23 0.22 0.21 0.20 VS = 3V
05614-032
-40 -60 -6
0.19 0.18
VS = 5V
-4
-2
0 VOUT (V)
2
4
6
Figure 18. Input Error Voltage vs. Output Voltage
Figure 21. Small Signal Transient Response for Various Supplies
Rev. A | Page 9 of 20
05614-031
1
05614-013
0.10
20pF 47pF
ADA4841-1
6 VIN 5 VOUT VS = 5V G = +1 TIME = 200ns/DIV 0.130 0.125 0.120 G = +1 TIME = 50ns/DIV VS = 3V
INPUT AND OUTPUT VOLTAGE (V)
OUTPUT VOLTAGE (V)
4
0.115 0.110 0.105 0.100
VS = 5V
3 2
1
05614-019
-1
0.090
Figure 22. Input Overdrive Recovery
6
Figure 25. Small Signal Transient Response for Various Supplies
4.5 4.0 3.5
VIN x 2 VOUT
INPUT AND OUTPUT VOLTAGE (V)
5 4 3 2 1 0 -1
VS = 5V G = +2 TIME = 100ns/DIV
G = +2 VS = 5 TIME = 100ns/DIV
OUTPUT VOLTAGE (V)
+125C 3.0 2.5 -40C 2.0 1.5 1.0
+25C
05614-023
0.5 0
Figure 23. Output Overdrive Recovery
Figure 26. Output Voltage vs. Temperature
1.5
1.0
VS = 5V VOUT = 2V p-p TIME = 100ns/DIV
2.0 1.5 1.0 VOUT
G = +1
VS = 5V G = +1 VOUT = 2V p-p TIME = 100ns/DIV
2.0 1.5 1.0
OUTPUT VOLTAGE (V)
G = +2 0.5
0
0 -0.5 -1.0
VIN
VOUT (EXPANDED) 0 -0.5 -1.0
-0.5
-1.0
05614-022
-1.5
-2.0
-2.0
Figure 24. Large Signal Transient Response for Various Gains
Figure 27. Settling Time
Rev. A | Page 10 of 20
05614-041
-1.5
-1.5
VIN, VOUT (V)
VOUT (mV)
0.5
0.5
05614-016
05614-030
0
0.095
ADA4841-1
-40
6 POWER DOWN PIN 5 ENA +25C 1.0 0.8 ENA -40C 3 ENA +125C 0.6 VOUT (V) 1.2
VS = 5V G = +2
COMMON-MODE REJECTION (dB)
-50 -60 -70 -80 -90 -100
POWER DOWN PIN (V)
4
2
0.4
1 0 -1 VS = 5V G = +1 VIN = 1VDC TIME = 200ns/DIV
0.2
05614-039
-0.2
-110 100
1k
10k
100k
1M
10M
100M
FREQUENCY (Hz)
Figure 28. Power-Up Time vs. Temperature
Figure 31. CMRR vs. Frequency
0
6 POWER DOWN PIN 5 VS = 5V G = +1 VIN = 1VDC 1.0 TIME = 10s/DIV 0.8
VOUT (V)
1.2
VS = 5V
POWER SUPPLY REJECTION (dB)
-20
POWER DOWN PIN (V)
4
-40
+PSR
3 +125C 2 +25C -40C
0.6
-60
0.4
-80 -PSR -100
05614-025
1
0.2
-1
POWER DOWN PIN
05614-040
0
0 -0.2
-120 100
1k
10k
100k
1M
10M
100M
FREQUENCY (Hz)
Figure 29. Power-Down Time vs. Temperature
Figure 32. PSRR vs. Frequency
1.6 VS = 5V 1.4 1.2 +125C
CLOSED-LOOP OUTPUT IMPEDANCE ()
100
VS = 5V
10
SUPPLY CURRENT (mA)
1.0 0.8 0.6 0.4 0.2 0 -0.2 0 0.5 1.0 1.5 2.0 2.5 3.0
+25C
1
-40C
0.1
0.01
05614-024
05614-020
3.5
4.0
4.5
5.0
0.001 100
1k
10k
100k
1M
10M
100M
POWER DOWN PIN (V)
FREQUENCY (Hz)
Figure 30. Supply Current vs. POWER DOWN Pin Voltage
Figure 33. Output Impedance vs. Frequency
Rev. A | Page 11 of 20
05614-009
0
ADA4841-1
40 30
INPUT OFFSET VOLTAGE (V)
1.6 1.5 VS = +5V
SUPPLY CURRENT (mA)
20 10 0 VS = 5V -10 -20 -30
1.4 1.3 1.2 1.1 1.0 VS = +5V
05614-059
VS = 5V
VS = +3V
-40 -50 -40
05614-057
0.9 VS = +3V 0.8 -40 -25 -10 5 20 35 50 65 80 95 110
-25
-10
5
20
35
50
65
80
95
110
125
125
TEMPERATURE (C)
TEMPERATURE (C)
Figure 34. Input Offset Voltage vs. Temperature for Various Supplies
Figure 36. Supply Current vs. Temperature for Various Supplies
3.6
3.5
INPUT BIAS CURRENT (A)
VS = +5V 3.4
3.3
VS = +3V VS = 5V
3.2
05614-058
3.1 -40
-25
-10
5
20
35
50
65
80
95
110
125
TEMPERATURE (C)
Figure 35. Input Bias Current vs. Temperature for Various Supplies
Rev. A | Page 12 of 20
ADA4841-1 THEORY OF OPERATION
AMPLIFIER DESCRIPTION
The ADA4841-1 is a low power, low noise, precision voltagefeedback op amp for single or dual voltage supply operation. The ADA4841-1 is fabricated on ADI's second generation XFCB process and features a trimmed supply current and an offset voltage. The 2 nV/Hz voltage noise (very low for a 1.1 mA supply current amplifier), 60 V offset voltage, and sub 1 V/C offset drift is accomplished with an input stage made of an undegenerated PNP input pair driving a symmetrical folded cascode. A rail-to-rail output stage provides the maximum linear signal range possible on low voltage supplies and has the current drive capability needed for the relatively low resistance feedback networks required for low noise operation. CMRR, PSRR, and open-loop gain are all well above 100 dB, preserving the precision performance in a variety of configurations. Gain bandwidth is kept high for this power level to preserve the outstanding linearity performance to frequencies up to 100 kHz. The ADA4841-1 has a power-down function to further reduce power consumption. All this results in a low noise, power efficient, precision amplifier that is well suited for high resolution and precision applications.
The total output voltage error is the sum of errors due to the amplifier's offset voltage and input currents. The output error due to the offset voltage can be estimated as
VOUTERROR = VCM VP - VPNOM VOUT RF + + VOFFSETNOM + x 1 + CMRR PSRR A RG where: VOFFSETNOM is the offset voltage at the specified supply voltage. This is measured with the input and output at midsupply. VCM is the common-mode voltage. VP is the power supply voltage. VpNOM is the specified power supply voltage. CMRR is the common-mode rejection ratio. PSRR is the power supply rejection ratio. A is the dc open-loop gain. The output error due to the input currents can be estimated as R VOUTERROR = (RF || RG ) x 1 + F R G R I B - - RS x 1 + F R G x I B+ (5) (4)
DC PERFORMANCE CONSIDERATIONS
Figure 37 shows a typical connection diagram and the major dc error sources. The ideal transfer function (all error sources set to 0 and infinite dc gain) can be written as
R VOUT = 1 + F R G x VIP R - F R G
RF - VIN + RG IB- - VIP + RS
05614-004
x VIN
(1)
Note that setting RS equal to RF||RG compensates for the voltage error due to the input bias current.
+ VOS - + VOUT -
NOISE CONSIDERATIONS
Figure 38 illustrates the primary noise contributors for the typical gain configurations. The total rms output noise is the root-mean-square of all the contributions.
RF vn _ RG = 4kT x RG RG ien ven + vout_en - vn _ RF = 4kT x RF
IB+
Figure 37. Typical Connection Diagram and DC Error Sources
This reduces to the familiar forms for inverting and noninverting op amp gain expressions:
vn _ RS = 4kT x RS
RS ien
05614-005
R VOUT = 1 + F R G
xVIP
(2)
Figure 38. Noise Sources in Typical Connection
(Noninverting gain, VIN = 0 V)
- RF VOUT = R G xVIN
(3)
(Inverting gain, VIP = 0 V)
Rev. A | Page 13 of 20
ADA4841-1
The output noise spectral density can be calculated by
vout _ en = R R 2 2 4 kTRf + 1 + F 4 kTRs + ien RS 2 + ven + F 4 kTRg + ien 2 R F 2 R R G G
2
[
]
2
(6) where: k is Boltzmann's Constant. T is the absolute temperature, degrees Kelvin. ien is the amplifier input current noise spectral density, pA/Hz.
The input stage positive limit is almost exactly a volt below the positive supply at room temperature. Input voltages above that start to show clipping behavior. The positive input voltage limit increases with temperature with a coefficient of about 2 mV/C. The lower supply limit is nominally below the minus supply; therefore, in a standard gain configuration, the output stage limits the signal headroom on the negative supply side. Figure 40 and Figure 41 show the nominal CMRR behavior at the limits of the input headroom for three temperatures--this is generated using the subtractor topology shown in Figure 42, which avoids the output stage limitation.
300 260 220 180 140 100 60 20 -20 -60 -100 -140 -180 -260 -300 3.00 3.20 3.40 3.60 3.80 4.00 4.20 4.40 4.60
05614-055
COMMON-MODE ERROR (V)
ven is the amplifier input voltage spectral density, nV/Hz. RS is the source resistance as shown in Figure 38. RF and RG are the feedback network resistances, as shown in Figure 38. Source resistance noise, amplifier voltage noise (ven), and the voltage noise from the amplifier's current noise (ien x RS) are all subject to the noise gain term (1 + RF/RG). Note that with a 2 nV/Hz input voltage noise and 1 pA/Hz input current, the noise contributions of the amplifier are relatively small for source resistances between approximately 200 and 30 k. Figure 39 shows the total RTI noise due to the amplifier vs. the source resistance. In addition, the value of the feedback resistors used impacts the noise. It is recommended to keep the value of feedback resistors between 250 and 1 k to keep the total noise low.
1000
+125C +25C -40C
-220
4.80 5.00
COMMON-MODE VOLTAGE (V)
Figure 40. VOS vs. +CMP vs. Common-Mode Error
0 -50 -100
COMMON-MODE ERROR (V)
-150 -200 -250 -300 -350 -400 -450 -500 -550 -600 -650
-40C +25C
100
NOISE (nV/ Hz)
AMPLIFIER + RESISTOR NOISE
+125C
10
TOTAL AMPLIFIER NOISE 1
05614-007
-700 -750 -800 -6.00 -5.80 -5.60 -5.40 -5.20 -5.00 -4.80 -4.60 -4.40 -4.20 -4.00 COMMON-MODE VOLTAGE (V)
SOURCE RESISTANCE NOISE 0.1 10 100 1k SOURCE RESISTANCE () 10k
Figure 41. VOS vs. -CMP vs. Common-Mode Error
- VCM + + VOUT -
100k
Figure 39. RTI Noise vs. Source Resistance
HEADROOM CONSIDERATIONS
The ADA4841-1 is designed to provide maximum input and output signal ranges with 16-bit to 18-bit dc linearity. As the input or output headroom limits are reached, the signal's linearity degrades.
Rev. A | Page 14 of 20
Figure 42. Common-Range Subtractor
05614-051
05614-054
ADA4841-1
Figure 43 shows the amplifier's frequency response as a G = -1 inverter with the input and output stage biased near the negative supply rail.
6 VS+ = 5V G = -1 VIN = 20mV p-p
60 G = +1 50
SERIES RESISTANCE ()
VS- = -150mV VS- = -100mV VS- = -200mV
40
3
VS- = -50mV 0
30
GAIN (dB)
20
-3
VS- = -20mV
10
-6
G = +5 0 10 100 1000
10000
-9
05614-017
CAPACITANCE LOAD (pF)
Figure 44. Series Resistance vs. Capacitive Load
-12 0.1
1
10 FREQUENCY (MHz)
100
INPUT PROTECTION
The ADA4841-1 is fully protected from ESD events, withstanding ESD events of 2.5 keV with no measured performance degradation. The precision input is protected with an ESD network between the power supplies and diode clamps across the input device pair, as shown in Figure 45.
VCC BIAS ESD VP ESD VN ESD
Figure 43. Small Signal Frequency Response vs. Negative Supply Bias
The input voltage (VIN) and reference voltage (VIP) are both at 0 V, see Figure 37. +VS is biased at +5 V, and -VS is swept from -200 mV to -20 mV. With the input and output voltages biased 200 mV from the bottom rail, the G = -1 inverter frequency response is not much different from what is seen with the input and output voltages biased near midsupply. At 150 mV bias, the frequency response starts to decrease and at 20 mV, the inverter band-width is less than half its nominal value.
CAPACITANCE DRIVE
Capacitance at the output of an amplifier creates a delay within the feedback path that, if within the bandwidth of the loop, can create excessive ringing and oscillation. The G = +1 follower topology has the highest loop bandwidth of any typical configuration and therefore is the most vulnerable to the effects of capacitance load. A small resistor in series with the amplifier's output and the capacitive load mitigates the problem. Figure 44 plots the recommended series resistance vs. the capacitance for gains of +1, +2, and +5.
ESD VEE
TO REST OF AMPLIFIER
Figure 45. Input Stage and Protection Diodes
For differential voltages above approximately 1.4 V, the diode clamps start to conduct. Too much current can cause damage due to excessive heating. If large differential voltages need to be sustained across the input terminals, it is recommended that the current through the input clamps be limited to below 150 mA. Series input resistors sized appropriately for the expected differential overvoltage provide the needed protection. The ESD clamps start to conduct for input voltages more than 0.7 V above the positive supply and input voltages more than 0.7 V below the negative supply. It is recommended that the fault current be limited to less than 150 mA if an overvoltage condition is expected.
Rev. A | Page 15 of 20
05614-006
05614-050
G = +2
ADA4841-1
POWER-DOWN OPERATION
Figure 46 shows the ADA4841-1 power-down circuitry. If the POWER DOWN pin is left unconnected, the base of Qx is pulled high by resistor Y and the part is turned on. Pulling the power-down pin approximately 1.7 V below the positive supply turns the part off, reducing the supply current to approximately 40 A.
VCC IBIAS ESD POWER DOWN ESD TO AMPLIFIER BIAS
VEE
Figure 46. Power-Down Circuit
The POWER DOWN pin is protected with ESD clamps, as shown in Figure 46. Voltages beyond the power supplies cause these diodes to conduct. The guidelines for limiting the overload current in the input protection section should also be followed for the POWER DOWN pin.
05614-052
Rev. A | Page 16 of 20
ADA4841-1 APPLICATIONS
TYPICAL PERFORMANCE VALUES
To reduce design time and eliminate uncertainty Table 6 provides a convenient reference for typical gains, component values, and performance parameters.
RECONSTRUCTION FILTER
The ADA4841-1 can also be used as a reconstruction filter at the output of DACs for suppression of the sampling frequency. The filter shown in Figure 48 is a two-pole, 500 KHz Sallen-Key LPF with a fixed gain of G = +1.6.
C2 660pF +5V 10F
16-BIT ADC DRIVER
The combination of low noise, low power, and high speed make the ADA4841-1 the perfect driver solution for low power, 16-bit ADCs, such as the AD7685. Figure 47 shows a typical 16-bit single-supply application. There are different challenges to do a single-supply, high resolution design, and the ADA4841-1 addresses these nicely. In a single-supply system, one of the main challenges is to use the amplifier in buffer mode to have the lowest output noise and still preserve linearity compatible with the ADC. Rail-to-rail input amplifiers are usually higher noise than the ADA4841-1 and cannot be used in this mode because of the nonlinear region around the crossover point of their input stages. The ADA4841-1, which has no crossover region but has a wide linear input range from 100 mV below ground to 1 V below positive rail, solves this problem, as shown in Figure 47. It can accept the 0 V to 4.096 V input range with a supply as low as 5.2 V. This supply also allows the use of a small, low dropout, low temperature drift ADR364 reference voltage. Note that at the low end of the input range close to ground, the ADA4841-1 can exhibit some nonlinearity, such as any rail-to-rail output amplifier. The ADA4841-1 drives a one-pole, low-pass filter. This filter limits the already very low noise contribution from the amplifier to the AD7685.
+5.2V
0.1F INPUT RT 49 R1 499 R2 499 C1 660pF
U1
0.1F
OUTPUT
10F -5V R3 499 R4 280
05614-044
Figure 48. Two-Pole 500 kHz Reconstruction Filter Schematic
Setting the resistors and capacitors equal to each other greatly simplifies the design equations for the Sallen-Key filter. The corner frequency, or -3 dB frequency, can be described by the equation
fC = 1 2R1C1
The quality factor, or Q, is shown in the equation
Q= 1 3-K
ADR364 100nF 10F
100nF
For minimum peaking, set Q equal to 0.707. The gain, or K, of the amplifier is
ADA4841 0V TO 4.096V
100nF 33 REF IN+ VDD VIO SDI SCK
K=
R4 +1 R3
2.7nF IN- GND
AD7685
SDO CNV
05614-060
Resistor values are kept low for minimal noise contribution, offset voltage, and optimal frequency response.
Figure 47. ADC Driver Schematic
Table 6. Recommended Values and Typical Performance
Gain +1 +2 -1 +5 +10 +20 RF () 0 499 499 499 499 499 RG () N/A 499 499 124 54.9 26.1 -3 dB BW (MHz) 77 34 38 11 5 2.3 Slew Rate (V/s) 12.5 12.5 12.5 12 12 11.2 Peaking (dB) 0.9 0.3 0.4 0 0 0 Output Noise ADA4841-1 Only (nV/Hz) 2 4 4 10 20 40 Total Output Noise Including Resistors (nV/Hz) 2 5.73 5.73 11.9 21.1 42.2
Rev. A | Page 17 of 20
ADA4841-1
Capacitor selection is critical for optimal filter performance. Capacitors with low temperature coefficients, such as NPO ceramic capacitors, are good choices for filter elements. Figure 49 shows the filter response.
5 0 -5 -10
GAIN (dB)
POWER SUPPLY BYPASSING
Power supply bypassing is a critical aspect in the performance of the ADA4841-1. A parallel connection of capacitors from each of the power supply pins to ground works best. A typical connection is shown in Figure 48. Smaller value capacitors offer better high frequency response where larger value electrolytics offer better low frequency performance. Paralleling different values and sizes of capacitors helps to ensure that the power supply pins are provided a low ac impedance across a wide band of frequencies. This is important for minimizing the coupling of noise into the amplifier. This can be especially important when the amplifier PSR is starting to roll off--the bypass capacitors can help lessen the degradation in PSR performance. Starting directly at the ADA4841-1 power supply pins, the smallest value capacitor should be placed on the same side of the board as the amplifier, and as close as possible to the amplifier power supply pin. The ground end of the capacitor should be connected directly to the ground plane. Keeping the capacitors' distance short but equal from the load is important and can improve distortion performance. This process should be repeated for the next largest value capacitor. It is recommended that a 0.01 F ceramic 0508 case be used. The 0508 case size offers low series inductance and excellent high frequency performance. A 10 F electrolytic capacitor should be placed in parallel with the 0.01 F capacitor. Depending on the circuit parameters, some enhancement to performance can be realized by adding additional capacitors. Each circuit is different and should be individually analyzed for optimal performance.
-15 -20 -25 -30 -35 -40 0.03
05614-043
0.1
1 FREQUENCY (MHz)
10
Figure 49. Filter Frequency Response
LAYOUT CONSIDERATIONS
To ensure optimal performance, careful and deliberate attention must be paid to the board layout, signal routing, power supply bypassing, and grounding.
GROUND PLANE
It is important to avoid ground in the areas under and around the input and output of the ADA4841-1. Stray capacitance created between the ground plane and the input and output pads of a device are detrimental to a high speed amplifier's performance. Stray capacitance at the inverting input, along with the amplifier's input capacitance, lowers the phase margin and can cause instability. Stray capacitance at the output creates a pole in the feedback loop. This can reduce phase margin and can cause the circuit to become unstable.
Rev. A | Page 18 of 20
ADA4841-1 OUTLINE DIMENSIONS
5.00 (0.1968) 4.80 (0.1890)
8 5 4
4.00 (0.1574) 3.80 (0.1497) 1
6.20 (0.2440) 5.80 (0.2284)
1.27 (0.0500) BSC 0.25 (0.0098) 0.10 (0.0040)
1.75 (0.0688) 1.35 (0.0532)
0.50 (0.0196) x 45 0.25 (0.0099)
0.51 (0.0201) COPLANARITY SEATING 0.31 (0.0122) 0.10 PLANE
8 0.25 (0.0098) 0 1.27 (0.0500) 0.40 (0.0157) 0.17 (0.0067)
COMPLIANT TO JEDEC STANDARDS MS-012-AA CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS (IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN
Figure 50. 8-Lead Standard Small Outline Package [SOIC_N] Narrow Body (R-8) Dimensions shown in millimeters and (inches)
ORDERING GUIDE
Model ADA4841-1YRZ 1 ADA4841-1YRZ-R71 ADA4841-1YRZ-RL1
1
Temperature Range -40C to +125C -40C to +125C -40C to +125C
Package Description 8-Lead SOIC_N 8-Lead SOIC_N 8-Lead SOIC_N
Package Option R-8 R-8 R-8
Ordering Quantity 1 1,000 2,500
Z = Pb-free part.
Rev. A | Page 19 of 20
ADA4841-1 NOTES
(c) 2005 Analog Devices, Inc. All rights reserved. Trademarks and registered trademarks are the property of their respective owners. D05614-0-9/05(A)
Rev. A | Page 20 of 20

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